View AD743_20915.PDF datasheet online --- IC-ON-LINE

Datasheet File OCR Text:

a
FEATURES ULTRALOW NOISE PERFORMANCE 2.9 nV/Hz at 10 kHz 0.38 V p-p, 0.1 Hz to 10 Hz 6.9 fA/Hz Current Noise at 1 kHz EXCELLENT DC PERFORMANCE 0.5 mV max Offset Voltage 250 pA max Input Bias Current 1000 V/mV min Open-Loop Gain AC PERFORMANCE 2.8 V/ s Slew Rate 4.5 MHz Unity-Gain Bandwidth THD = 0.0003% @ 1 kHz Available in Tape and Reel in Accordance with EIA-481A Standard APPLICATIONS Sonar Preamplifiers High Dynamic Range Filters (>140 dB) Photodiode and IR Detector Amplifiers Accelerometers PRODUCT DESCRIPTION 8-Pin Plastic Mini-DIP (N) and 8-Pin Cerdip (Q) Packages
NULL -IN +IN -V S 1 2 3 4 TOP VIEW
Ultralow Noise BiFET Op Amp AD743
CONNECTION DIAGRAMS 16-Pin SOIC (R) Package
NC 1 2 3 4 5 6 7 8 8 16 NC NC
AD743
8 7 6 5
NC
OFFSET +V S NULL OUT NULL -IN NC +IN
AD743
15
14 NC 13 12 11 10 9 +VS OUTPUT OFFSET NULL NC NC
NC = NO CONNECT
-V S NC NC
NC = NO CONNECT
PRODUCT HIGHLIGHTS
1. The low offset voltage and low input offset voltage drift of the AD743 coupled with its ultralow noise performance mean that the AD743 can be used for upgrading many applications now using bipolar amplifiers. 2. The combination of low voltage and low current noise make the AD743 ideal for charge sensitive applications such as accelerometers and hydrophones. 3. The low input offset voltage and low noise level of the AD743 provide >140 dB dynamic range. 4. The typical 10 kHz noise level of 2.9 nV/Hz permits a three op amp instrumentation amplifier, using three AD743s, to be built which exhibits less than 4.2 nV/Hz noise at 10 kHz and which has low input bias currents.
1000 R SOURCE EO R SOURCE OP27 & RESISTOR (--)
The AD743 is an ultralow noise precision, FET input, monolithic operational amplifier. It offers a combination of the ultralow voltage noise generally associated with bipolar input op amps and the very low input current of a FET-input device. Furthermore, the AD743 does not exhibit an output phase reversal when the negative common-mode voltage limit is exceeded. The AD743's guaranteed, maximum input voltage noise of 4.0 nV/Hz at 10 kHz is unsurpassed for a FET-input monolithic op amp, as is the maximum 1.0 V p-p, 0.1 Hz to 10 Hz noise. The AD743 also has excellent dc performance with 250 pA maximum input bias current and 0.5 mV maximum offset voltage. The AD743 is specifically designed for use as a preamp in capacitive sensors, such as ceramic hydrophones. It is available in five performance grades. The AD743J and AD743K are rated over the commercial temperature range of 0C to +70C. The AD743A and AD743B are rated over the industrial temperature range of -40C to +85C. The AD743S is rated over the military temperature range of -55C to +125C and is available processed to MIL-STD-883B, Rev. C. The AD743 is available in 8-pin plastic mini-DIP, 8-pin cerdip, 16-pin SOIC, or in chip form.
INPUT NOISE VOLTAGE - nV/ Hz
100
AD743 & RESISTOR OR OP27 & RESISTOR 10
AD743 + RESISTOR ) (
RESISTOR NOISE ONLY (- - -) 1 100
1k
10k
100k
1M
10M
SOURCE RESISTANCE -
Input Noise Voltage vs. Source Resistance
REV. C
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703
AD743-SPECIFICATIONS (@ +25 C and
Model INPUT OFFSET VOLTAGE1 Initial Offset Initial Offset vs. Temp. vs. Supply (PSRR) vs. Supply (PSRR) INPUT BIAS CURRENT3 Either Input Either Input @ TMAX Either Input Either Input, VS = 5 V INPUT OFFSET CURRENT Offset Current @ TMAX FREQUENCY RESPONSE Gain BW, Small Signal Full Power Response Slew Rate, Unity Gain Settling Time to 0.01% Total Harmonic Distortion4 (Figure 16) INPUT IMPEDANCE Differential Common Mode INPUT VOLTAGE RANGE Differential5 Common-Mode Voltage Over Max Operating Range6 Common-Mode Rejection Ratio INPUT VOLTAGE NOISE Conditions Min AD743J Typ 0.25 TMIN to TMAX TMIN to TMAX 12 V to 18 V2 TMIN to TMAX VCM = 0 V VCM = 0 V VCM = +10 V VCM = 0 V VCM = 0 V VCM = 0 V G = -1 VO = 20 V p-p G = -1 f = 1 kHz G = -1 4.5 25 2.8 6 0.0003 1 3 1010||20 1011||18 2 96
15 V dc, unless otherwise noted)
Max 1.0/0.8 1.5 100 98 400 8.8/25.6 600 200 150 2.2/6.4 4.5 25 2.8 6 0.0003 1 3 1010||20 1011||18 Min AD743K/B Typ 0.1 1 106 100 150 250 5.5/16 400 125 75 1.1/3.2 4.5 25 2.8 6 0.0003 1 3 1010||20 1011||18 Max 0.5/0.25 1.0/0.50 90 88 Min AD743S Typ 0.25 2 96 Max 1.0 2.0 Units mV mV V/C dB dB pA nA pA pA pA nA MHz kHz V/s s % ||pF ||pF V V V dB dB V p-p nV/Hz nV/Hz nV/Hz nV/Hz fA/Hz V/mV V/mV V/mV V V V V mA V V mA
90 88
150
150
400 413 600 200 150 102
250 30 40
250 30 30
300 30 40
-10 VCM = 10 V TMIN to TMAX 0.1 Hz to 10 Hz f = 10 Hz f = 100 Hz f = 1 kHz f = 10 kHz f = 1 kHz VO = 10 V RLOAD 2 k TMIN to TMAX RLOAD = 600 1000 800 80 78
20 +13.3, -10.7 +12 95 0.38 5.5 3.6 3.2 2.9 6.9 4000 1200
-10 90 88
20 +13.3, -10.7 +12 102 0.38 5.5 3.6 3.2 2.9 6.9 1.0 10.0 6.0 5.0 4.0
-10 80 78
20 +13.3, -10.7 +12 95 0.38 5.5 3.6 3.2 2.9 6.9
5.0 4.0
5.0 4.0
INPUT CURRENT NOISE OPEN LOOP GAIN
2000 1800
4000 1200
1000 800
4000 1200
OUTPUT CHARACTERISTICS Voltage RLOAD 600 RLOAD 600 TMIN to TMAX RLOAD 2 k Current Short Circuit POWER SUPPLY Rated Performance Operating Range Quiescent Current TRANSISTOR COUNT # of Transistors
+13, -12 +13.6, -12.6 +12, -10 12 +13.8, -13.1 20 40 15 8.1 50
+13, -12 +13.6, -12.6 +12, -10 12 +13.8, -13.1 20 40 15 8.1 50
+13, -12 +13.6, -12.6 +12, -10 12 +13.8, -13.1 20 40 15 8.1 50
4.8
18 10.0
4.8
18 10.0
4.8
18 10.0
NOTES 1 Input offset voltage specifications are guaranteed after 5 minutes of operation at TA = +25C. 2 Test conditions: +VS = 15 V, -VS = 12 V to 18 V and +VS = 12 V to +18 V, -VS = 15 V. 3 Bias current specifications are guaranteed maximum at either input after 5 minutes of operation at TA = +25C. For higher temperature, the current doubles every 10C. 4 Gain = -1, RL = 2 k, CL = 10 pF. 5 Defined as voltage between inputs, such that neither exceeds 10 V from common. 6 Thc AD743 does not exhibit an output phase reversal when the negative common-mode limit is exceeded. All min and max specifications are guaranteed. Specifications subject to change without notice.
-2-
REV. C
AD743
ABSOLUTE MAXIMUM RATINGS 1
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V Internal Power Dissipation2 Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . VS Output Short Circuit Duration . . . . . . . . . . . . . . . . . Indefinite Differential Input Voltage . . . . . . . . . . . . . . . . . . +VS and -VS Storage Temperature Range (Q) . . . . . . . . . . -65C to +150C Storage Temperature Range (N, R) . . . . . . . . -65C to +125C Operating Temperature Range AD743J/K . . . . . . . . . . . . . . . . . . . . . . . . . . . 0C to +70C AD743A/B . . . . . . . . . . . . . . . . . . . . . . . . . . -40C to +85C AD743S . . . . . . . . . . . . . . . . . . . . . . . . . . . -55C to +125C Lead Temperature Range (Soldering 60 seconds) . . . . . 300C
NOTES 1 Stresses above those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 8-pin plastic package: JA = 100C/Watt, JC = 50C/Watt 8-pin cerdip package: JA = 110C/Watt, JC = 30C/Watt 16-pin plastic SOIC package: JA = 100C/Watt, JC = 30C/Watt
ORDERING GUIDE
Model AD743JN AD743KN AD743JR-16 AD743KR-16 AD743BQ AD743SQ/883B AD743JR-16-REEL AD743KR-16-REEL
Temperature Range 0C to +70C 0C to +70C 0C to +70C 0C to +70C -40C to +85C -55C to +125C 0C to +70C 0C to +70C
Package Option* N-8 N-8 R-16 R-16 Q-8 Q-8 Tape & Reel Tape & Reel
*N = Plastic DIP; R = Small Outline IC; Q = Cerdip.
ESD SUSCEPTIBILITY
An ESD classification per method 3015.6 of MIL-STD-883C has been performed on the AD743. The AD743 is a class 1 device, passing at 1000 V and failing at 1500 V on null pins 1 and 5, when tested, using an IMCS 5000 automated ESD tester. Pins other than null pins fail at greater than 2500 V.
METALIZATION PHOTOGRAPH
Contact factory for latest dimensions. Dimensions shown in inches and (mm).
REV. C
-3-
AD743 -Typical Characteristics (@ +25 C, V = +15 V)
S
20 R LOAD = 10k 15
20 R LOAD = 10k
OUTPUT VOLTAGE SWING - Volts p-p
35
30
OUTPUT VOLTAGE SWING - Volts
INPUT VOLTAGE SWING - Volts
15 POSITIVE SUPPLY
+VIN
25
20
10
10 NEGATIVE SUPPLY 5
15
-VIN
5
10
5 0
0 0 5 10 15 20 SUPPLY VOLTAGE VOLTS
0 0 5 15 SUPPLY VOLTAGE VOLTS 10 20
10
100 1k LOAD RESISTANCE -
10k
Figure 1. Input Voltage Swing vs. Supply Voltage
12
Figure 2. Output Voltage Swing vs. Supply Voltage
-6
Figure 3. Output Voltage Swing vs. Load Resistance
200 100
10
10
QUIESCENT CURRENT- mA
-7
OUTPUT IMPEDANCE -
9
INPUT BIAS CURRENT - Amps
10
-8
10
6
10
-9
1
10
-10
3
0.1
10
-11
0 0 5 10 15 20 SUPPLY VOLTAGE VOLTS
10
-12
0.01
-60 -40
-20
0
20
40
60
80 100
120
140
10k
100k
1M
10M
100M
TEMPERATURE - C
FREQUENCY - Hz
Figure 4. Quiescent Current vs. Supply Voltage
300
Figure 5. Input Bias Current vs. Temperature
80 70
Figure 6. Output Impedance vs. Frequency (Closed Loop Gain = -1)
7.0
60 200
CURRENT LIMIT - mA
+ OUTPUT CURRENT
50 40 30 20 10
GAIN BANDWIDTH PRODUCT - MHz
6.0
INPUT BIAS CURRENT - pA
5.0
4.0
100
- OUTPUT CURRENT
3.0
0 -12
0 -9 -6 -3 0 3 6 9 12 -60 -40 -20 0 20 40 60 80 100 120 140 COMMON MODE VOLTAGE - Volts TEMPERATURE - C
2.0 -60 -40
-20
0
20
40
60
80 100 120
140
TEMPERATURE - C
Figure 7. Input Bias Current vs. Common-Mode Voltage
Figure 8. Short Circuit Current Limit vs. Temperature
Figure 9. Gain Bandwidth Product vs. Temperature
-4-
REV. C
AD743
100 100
3.5
150
80 PHASE
80
140
PHASE MARGIN - Degrees 60
OPEN-LOOP GAIN - dB
60 GAIN 40
SLEW RATE - Volts/s
3.0
OPEN-LOOP GAIN - dB
130
40
20
20
120
2.5
100
0
0
-20 100
1k
1M 10k 100k FREQUENCY - Hz
10M
-20 100M
2.0 -60 -40
80
-20
0 20 40 60 80 100 120 140 TEMPERATURE - C
0
5
10
15
20
SUPPLY VOLTAGE VOLTS
Figure 10. Open-Loop Gain and Phase vs. Frequency
120
120
Figure 11. Slew Rate vs. Temperature (Gain = -1)
Figure 12. Open-Loop Gain vs. Supply Voltage, RLOAD = 2K
35
COMMON-MODE REJECTION - dB
100 VCM = 10V 80
POWER SUPPLY REJECTION - dB
100 + SUPPLY 80
30
OUTPUT VOLTAGE - Volts p-p
25
20
60
60
15 R L= 2k 10
40
40 - SUPPLY 20
20
5
0 100
1k
10k 100k FREQUENCY - Hz
1M
0 100
1k
10k
100k
1M
10M
100M
0 1k 10k 100k 1M FREQUENCY - Hz
FREQUENCY - Hz
Figure 13. Common-Mode Rejection vs. Frequency
-70
Figure 14. Power Supply Rejection vs. Frequency
100
NOISE VOLTAGE (REFERRED TO INPUT) - nV Hz
Figure 15. Large Signal Frequency Response
1k
-80
CLOSED-LOOP GAIN = 1
-90
CURRENT NOISE SPECTRAL DENSITY - fA/ Hz
10
100
THD - dB
-100 GAIN = +10
-110
-120 GAIN = -1 -130
1.0
CLOSED-LOOP GAIN = 10
10
-140 10
100
1k FREQUENCY - Hz
10k
100k
0.1 1 10 100 1k 10k 100k FREQUENCY - Hz 1M 10M
1.0 1 10 1k 100 FREQUENCY - Hz 10k 100k
Figure 16. Total Harmonic Distortion vs. Frequency
Figure 17. Input Noise Voltage Spectral Density
Figure 18. Input Noise Current Spectral Density
REV. C
-5-
AD743 -Typical Characteristics (@ +25C, V = +15 V)
S
69 63 57
NUMBER OF UNITS
51 45 39 33 27 21 15 9 3 2.5 2.7 3.3 2.9 3.1 INPUT VOLTAGE NOISE - nV Hz 3.5 3.8
Figure 22b. Unity-Gain Follower Small Signal Pulse Response
100pF 2k +V S 2k VIN 7 2 1F 0.1F
Figure 19. Typical Noise Distribution @ 10 kHz (602 Units)
AD743
3 SQUARE WAVE INPUT 4 -VS 1F
6
VOUT CL 100pF
0.1F
Figure 23a. Unity-Gain Inverter Figure 20. Offset Null Configuration
Figure 21. Unity-Gain Follower
Figure 23b. Unity-Gain Inverter Large Signal Pulse Response
Figure 22a. Unity-Gain Follower Large Signal Pulse Response
Figure 23c. Unity-Gain Inverter Small Signal Pulse Response
-6-
REV. C
AD743
OP AMP PERFORMANCE: JFET VS. BIPOLAR DESIGNING CIRCUITS FOR LOW NOISE
The AD743 is the first monolithic JFET op amp to offer the low input voltage noise of an industry-standard bipolar op amp without its inherent input current errors. This is demonstrated in Figure 24, which compares input voltage noise vs. input source resistance of the OP27 and the AD743 op amps. From this figure, it is clear that at high source impedance the low current noise of the AD743 also provides lower total noise. It is also important to note that with the AD743 this noise reduction extends all the way down to low source impedances. The lower dc current errors of the AD743 also reduce errors due to offset and drift at high source impedances (Figure 25).
1000 R SOURCE EO R SOURCE OP27 & RESISTOR (--)
INPUT NOISE VOLTAGE - nV/ Hz
100
AD743 & RESISTOR OR OP27 & RESISTOR 10
AD743 + RESISTOR ) (
An op amp's input voltage noise performance is typicaly divided into two regions: flatband and low frequency noise. The AD743 offers excellent performance with respect to both. The figure of 2.9 nV/Hz @ 10 kHz is excellent for JFET input amplifier. The 0.1 Hz to 10 Hz noise is typically 0.38 V p-p. The user should pay careful attention to several design details in order to optimize low frequency noise performance. Random air currents can generate varying thermocouple voltages that appear as low frequency noise: therefore sensitive circuitry should be well shielded from air flow. Keeping absolute chip temperature low also reduces low frequency noise in two ways: first, the low frequency noise is strongly dependent on the ambient temperature and increases above +25C. Secondly, since the gradient of temperature from the IC package to ambient is greater, the noise generated by random air currents, as previously mentioned, will be larger in magnitude. Chip temperature can be reduced both by operation at reduced supply voltages and by the use of a suitable clip-on heat sink, if possible. Low frequency current noise can be computed from the ~ magnitude of the dc bias current ( In = 2qI B f ) and increases below approximately 100 Hz with a 1/f power spectral density. For the AD743 the typical value of current noise is 6.9 fA/Hz ~ at 1 kHz. Using the formula, In = 4kT /Rf , to compute the Johnson noise of a resistor, expressed as a current, one can see that the current noise of the AD743 is equivalent to that of a 3.45 108 source resistance. At high frequencies, the current noise of a FET increases proportionately to frequency. This noise is due to the "real" part of the gate input impedance, which decreases with frequency. This noise component usually is not important, since the voltage noise of the amplifier impressed upon its input capacitance is an apparent current noise of approximately the same magnitude. In any FET input amplifier, the current noise of the internal bias circuitry can be coupled externally via the gate-to-source capacitances and appears as input current noise. This noise is totally correlated at the inputs, so source impedance matching will tend to cancel out its effect. Both input resistance and input capacitance should be balanced whenever dealing with source capacitances of less than 300 pF in value.
LOW NOISE CHARGE AMPLIFIERS
RESISTOR NOISE ONLY (- - -) 1 100
1k
10k
100k
1M
10M
SOURCE RESISTANCE -
Figure 24. Total Input Noise Spectral Density @ 1 kHz vs. Source Resistance
100
INPUT OFFSET VOLTAGE - mV
ADOP27G 10
1.0
AD743 KN
0.1 100 1k 10k 100k 1M 10M SOURCE RESISTANCE -
Figure 25. Input Offset Voltage vs. Source Resistance
As stated, the AD743 provides both low voltage and low current noise. This combination makes this device particularly suitable in applications requiring very high charge sensitivity, such as capacitive accelerometers and hydrophones. When dealing with a high source capacitance, it is useful to consider the total input charge uncertainty as a measure of system noise. Charge (Q) is related to voltage and current by the simply stated fundamental relationships:
Q = CV and I = dQ dt
As shown, voltage, current and charge noise can all be directly related. The change in open circuit voltage (V) on a capacitor will equal the combination of the change in charge (Q/C) and the change in capacitance with a built in charge (Q/C).
REV. C
-7-
AD743
Figures 26 and 27 show two ways to buffer and amplify the output of a charge output transducer. Both require using an amplifier which has a very high input impedance, such as the AD743. Figure 26 shows a model of a charge amplifier circuit. Here, amplification depends on the principle of conservation of charge at the input of amplifier A1, which requires that the charge on capacitor CS be transferred to capacitor CF, thus yielding an output voltage of Q/CF. The amplifiers input voltage noise will appear at the output amplified by the noise gain (1 + (CS/CF)) of the circuit. Figure 28 shows that these two circuits have an identical frequency response and the same noise performance (provided that CS/CF = R1/ R2). One feature of the first circuit is that a "T" network is used to increase the effective resistance of RB and improve the low frequency cutoff point by the same factor.
-100 -110
DECIBELS REFERENCED TO 1V/ Hz
-120 -130 -140 -150 -160 -170 -180 -190 -200 -210 -220 10M 100M 1 10 100 FREQUENCY - Hz 1k 10k 100k
NOISE DUE TO R B ALONE NOISE DUE TO I B ALONE TOTAL OUTPUT NOISE
Figure 28. Noise at the Outputs of the Circuits of Figures 26 and 27. Gain = 10, CS = 3000 pF, RB = 22 M
Figure 26. A Charge Amplifier Circuit
However, this does not change the noise contribution of RB which, in this example, dominates at low frequencies. The graph of Figure 29 shows how to select an RB large enough to minimize this resistor's contribution to overall circuit noise. When the equivalent current noise of RB ((4kT)/R) equals the noise of IB ( 2qIB ), there is diminishing return in making RB larger.
5.2 x 10
10
RESISTANCE IN
5.2 x 10 9
5.2 x 10
8
Figure 27. Model for a High Z Follower with Gain
5.2 x 10
7
The second circuit, Figure 27, is simply a high impedance follower with gain. Here the noise gain (1 + (R1/R2)) is the same as the gain from the transducer to the output. Resistor RB, in both circuits, is required as a dc bias current return. There are three important sources of noise in these circuits. Amplifiers A1 and A2 contribute both voltage and current noise, while resistor RB contributes a current noise of:
T ~ f N = 4k RB
5.2 x 10 6 1pA 10pA 100pA 1nA INPUT BIAS CURRENT 10nA
Figure 29. Graph of Resistance vs. Input Bias Current where the Equivalent Noise 4kT/R, Equals the Noise of the Bias Current 2qIB
where: k = Boltzman's Constant = 1.381 x 10-23 Joules/Kelvin T = Absolute Temperature, Kelvin (0C = +273.2 Kelvin)
f = Bandwidth - in Hz (Assuming an Ideal "Brick Wall" Filter)
This must be root-sum-squared with the amplifier's own current noise.
To maximize dc performance over temperature, the source resistances should be balanced on each input of the amplifier. This is represented by the optional resistor RB in Figures 26 and 27. As previously mentioned, for best noise performance care should be taken to also balance the source capacitance designated by CB. The value for CB in Figure 26 would be equal to CS, in Figure 27. At values of CB over 300 pF, there is a diminishing impact on noise; capacitor CB can then be simply a large bypass of 0.01 F or greater.
-8-
REV. C
AD743
HOW CHIP PACKAGE TYPE AND POWER DISSIPATION AFFECT INPUT BIAS CURRENT
300
As with all JFET input amplifiers, the input bias current of the AD743 is a direct function of device junction temperature, IB approximately doubling every 10C. Figure 30 shows the relationship between bias current and junction temperature for the AD743. This graph shows that lowering the junction temperature will dramatically improve IB.
10
-6
TA = +25C 200 J A = 165C/W
100
J A = 115C/W
J A = 0C/W
10-7 VS = 15V TA = + 25C
0 5 10 SUPPLY VOLTAGE - Volts 15
10
-8
10-9
10
-10
Figure 32. Input Bias Current vs. Supply Voltage for Various Values of JA
10
-11
10
-12
-60 -40
-20 0 20 40 60 80 100 120 140 JUNCTION TEMPERATURE - C
Figure 30. Input Bias Current vs. Junction Temperature
The dc thermal properties of an IC can be closely approximated by using the simple model of Figure 31 where current represents power dissipation, voltage represents temperature, and resistors represent thermal resistance ( in C/Watt).
TJ JC JA CA
Figure 33. A Breakdown of Various Package Thermal Resistances
TA
P IN
REDUCED POWER SUPPLY OPERATION FOR LOWER IB
WHERE: PIN = DEVICE DISSIPATION TA = AMBIENT TEMPERATURE TJ = JUNCTION TEMPERATURE JC = THERMAL RESISTANCE - JUNCTION TO CASE CA = THERMAL RESISTANCE - CASE TO AMBIENT
Figure 31. A Device Thermal Model
Reduced power supply operation lowers IB in two ways: first, by lowering both the total power dissipation and second, by reducing the basic gate-to-junction leakage (Figure 32). Figure 34 shows a 40 dB gain piezoelectric transducer amplifier, which operates without an ac coupling capacitor, over the -40C to +85C temperature range. If the optional coupling capacitor is used, this circuit will operate over the entire -55C to +125C military temperature range.
From this model TJ = TA + JA Pin. Therefore, IB can be determined in a particular application by using Figure 30 together with the published data for JA and power dissipation. The user can modify JA by use of an appropriate clip-on heat sink such as the Aavid #5801. JA is also a variable when using the AD743 in chip form. Figure 32 shows bias current vs. supply voltage with JA as the third variable. This graph can be used to predict bias current after JA has been computed. Again bias current will double for every 10C. The designer using the AD743 in chip form (Figure 33) must also be concerned with both JC and CA, since JC can be affected by the type of die mount technology used. Typically, JC's will be in the 3C to 5C/watt range; therefore, for normal packages, this small power dissipation level may be ignored. But, with a large hybrid substrate, JC will dominate proportionately more of the total JA.
Figure 34. A Piezoelectric Transducer
AD743
AN INPUT-IMPEDANCE-COMPENSATED, SALLEN-KEY FILTER
The simple high pass filter of Figure 35 has an important source of error which is often overlooked. Even 5 pF of input capacitance in amplifier "A" will contribute an additional 1% of passband amplitude error, as well as distortion, proportional to the C/V characteristics of the input junction capacitance. The addition of the network designated "Z" will balance the source impedance-as seen by "A"-and thus eliminate these errors.
Figure 36b. An Accelerometer Circuit Employing a DC Servo Amplifier Figure 35. An Input Impedance Compensated Sallen-Key Filter
TWO HIGH PERFORMANCE ACCELEROMETER AMPLIFIERS
A dc servo-loop (Figure 36b) can be used to assure a dc output which is <10 mV, without the need for a large compensating resistor when dealing with bias currents as large as 100 nA. For optimal low frequency performance, the time constant of the servo loop (R4C2 = R5C3) should be:
R2 Time Constant 10 R11 + C1 R3
Two of the most popular charge-out transducers are hydrophones and accelerometers. Precision accelerometers are typically calibrated for a charge output (pC/g).* Figures 36a and 36b show two ways in which to configure the AD743 as a low noise charge amplifier for use with a wide variety of piezoelectric accelerometers. The input sensitivity of these circuits will be determined by the value of capacitor C1 and is equal to:
V OUT = QOUT C1
A LOW NOISE HYDROPHONE AMPLIFIER
The ratio of capacitor C1 to the internal capacitance (CT) of the transducer determines the noise gain of this circuit (1 + CT/C1). The amplifiers voltage noise will appear at its output amplified by this amount. The low frequency bandwidth of these circuits will be dependent on the value of resistor R1. If a "T" network is used, the effective value is: R1 (1 + R2/R3).
Hydrophones are usually calibrated in the voltage-out mode. The circuits of Figures 37a and 37b can be used to amplify the output of a typical hydrophone. Figure 37a shows a typical dc coupled circuit. The optional resistor and capacitor serve to counteract the dc offset caused by bias currents flowing through resistor R1. Figure 37b, a variation of the original circuit, has a low frequency cutoff determined by an RC time constant equal to:
Time Constant = 1 2 x CC x 100
Figure 36a. A Basic Accelerometer Circuit
*pC = Picocoulombs g = Earth's Gravitational Constant
Figure 37a. A Basic Hydrophone Amplifier
-10-
REV. C
AD743
Where the dc gain is 1 and the gain above the low frequency cutoff (1/(2CC(100 ))) is the same as the circuit of Figure 37a. The circuit of Figure 37c uses a dc servo loop to keep the dc output at 0 V and to maintain full dynamic range for IB's up to 100 nA. The time constant of R7 and C2 should be larger than that of R1 and CT for a smooth low frequency response. The transducer shown has a source capacitance of 7500 pF. For smaller transducer capacitances (300 pF), lowest noise can be achieved by adding a parallel RC network (R4 = R1, C1 = CT) in series with the inverting input of the AD743.
BALANCING SOURCE IMPEDANCES
Figure 37b. An AC-Coupled, Low Noise Hydrophone Amplifier
As mentioned previously, it is good practice to balance the source impedances (both resistive and reactive) as seen by the inputs of the AD743. Balancing the resistive components will optimize dc performance over temperature because balancing will mitigate the effects of any bias current errors. Balancing input capacitance will minimize ac response errors due to the amplifier's input capacitance and, as shown in Figure 38, noise performance will be optimized. Figure 39 shows the required external components for noninverting (A) and inverting (B) configurations.
Figure 38. RTI Voltage Noise vs. Input Capacitance Figure 37c. A Hydrophone Amplifier Incorporating a DC Servo Loop
Figure 39. Optional External Components for Balancing Source Impedances
REV. C
-11-
AD743
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
8-Pin Plastic Mini-DIP (N)
8-Pin Cerdip (Q) Packages
16-Pin SOIC (R) Package
-12-
REV. C
PRINTED IN U.S.A.
C1433-24-10/90

▲Up To Search▲

Price & Availability of AD743

	To Download AD743 Datasheet File
If you can't view the Datasheet, Please click here to try to view without PDF Reader .